It's possible to increase the accuracy of some systems by restricting the local temperature of certain crucial components to a narrow range.
THIERRY RAHBAN, Maxim Integrated Products
One way to stabilize component temperature is to enclose the component(s) within a fixed-temperature oven. To provide some margin for regulation, the chosen temperature should be higher than ambient under all conditions. This scheme is widely used, particularly in the design of extremely stable clocks such as oven-controlled crystal oscillators.
The need to regulate high-level temperature has some detrimental effects. First, the performance can be degraded slightly in several areas, including noise factor, speed, and lifetime. Second, the regulator consumes power for heating even when the ambient temperature lies in the middle of its range. Twice as much power is needed when the ambient temperature is at the lower end of its range. Third, the time required to reach a stable temperature can be fairly long, if the available electric power is limited.
A TEC consists of multiple semiconductor junctions connected electrically in series and bonded between two plates. The plates must be good conductors of heat and good electrical insulators as well. Ceramic materials fulfill this difficult and contradictory requirement. One plate is thermally connected to the ambient temperature, and the other is connected to the object whose temperature is to be regulated. Thanks to the Peltier Effect, a current through the junctions creates a temperature difference between the plates whose polarity and magnitude depend on the polarity and magnitude of the current. With respect to the ambient temperature, it then becomes possible to heat the object or cool it down. Today's technology allows temperature differences as high as 84°C, and cascading arrangements can produce even higher differences.
Temperature up, resistance down
A negative temperature coefficient (NTC) resistor is a temperature-sensitive device whose resistance decreases as its temperature increases. Among the many types of NTC components available, those fabricated with the ceramic-powder process exhibit the largest resistance change in response to minute variations of temperature. More important, some ceramic NTCs offer 0.05°C stability over their lifetimes-after proper aging. Compared with other temperature sensors, the size of ceramic NTCs can be surprisingly small.
Used in constant-temperature controllers, an NTC provides high sensitivity even when biased at today's very low supply voltages. The absolute error due to amplifier offset is close to 0.03°C for a 0.75-mV offset, while error due to self-heating of the NTC is 0.06°C in free air. (For sensors completely embedded in and enclosed by the material whose temperature is to be measured, the error is only half as much.) Fortunately, we are not concerned with absolute temperature error, but only with variation of this error within the working temperature range. Such variation is commonly an order of magnitude smaller than the absolute error.
Strategy of regulation
When subjected to heat flux, most systems (even small ones) exhibit an impressive delay before their temperature begins to stabilize. The time needed to attain 63.2% of a given temperature gradient is called the "thermal time constant," and it often ranges from 5 to 200 sec. Thus, the time constants associated with temperature-regulator operation can seem very long to an electronic engineer. The system discussed here has a thermal time constant of about 40 sec, which is slow indeed when compared with transients that are possible in the power-supply voltage.
To enable a response to any variation of the power-supply voltage, the design includes two parallel feedback paths (see Figure 1). One ceramic plate of the TEC is in close thermal contact with the object of interest (a laser diode in this case), and the other plate allows heat transfer to the external ambient temperature. This heat transfer should be as unimpeded as possible, and if necessary (when high power levels are encountered), it should be assisted with a blower. Because a certain level of continuous heat leakage is unavoidable, a corresponding amount of electric power is needed to compensate for the leakage at equilibrium.
To minimize error due to local drops in temperature, the point of temperature-sensing should be as close as possible to the object (the small footprint of an NTC is very convenient for that purpose). The measured temperature and the desired temperature are compared within a Wheatstone bridge. Amplifier A not only amplifies the error signal, but also provides the frequency phase and gain corrections necessary to stabilize the external closed loop. At any instant, it feeds the inner loop with the value of TEC current requested for attaining the proper temperature. This request is represented by a very slow signal, which can't react to fast variations in the power-supply voltage.
The inner loop regulates current into the TEC, and a switching regulator is mandatory to achieve the high-efficiency conversion that generates a minimum excess of heat. Because current ripple above 3% degrades the TEC's cooling efficiency, a high switching frequency is recommended for easy filtering of the AC components. The higher the frequency, the smaller the passive components. Inner-loop bandwidth must be sufficient to allow a response to the ripple and power-supply transients that resist normal filtering.
Each functional block of the controller is detailed.
H-bridge. The power stage must be able to feed the TEC with two polarities of current: one for cooling and the other for heating. For single-polarity power supplies, this goal is realized with an "H-bridge." When voltages at each leg of the H-bridge are equal (roughly at mid-supply voltage), the bridge is balanced and no current can circulate into the TEC. This principle works for linear and switching H-bridges as well.
Figure 2 illustrates the structure of the pulse-width modulation (PWM) H-bridge. The left leg of the bridge consists of two n-channel MOSFETs driven by the complementary signals DH and DL. To provide enough gate amplitude when the upper transistor is switched on, the DH signal is referenced to LX. DH at that time is about 3 V higher than LX, which is switched to the 3.3-V supply. Thus, the DH signal amplitude exceeds 6 V in its high state.
The DL signal, which requires no such boosting, switches between 0 and 3.3 V. Because MOSFETs in the right leg are driven in opposite phase with corresponding MOSFETs in the left leg, the DH signal now drives the lower transistor. The DL signal is not boosted to 6 V, so there is no way to use an n-MOSFET as the upper-left transistor. A p-channel transistor must be used, driven by the same DH signal as the lower transistor. To avoid any possibility of cross-conduction, the lower transistor is chosen for its high threshold (2.6-V minimum), and the upper transistor needs at least 0.85 V of gate drive to conduct. Thus, the two transistors cannot conduct simultaneously for supply voltages below 3.45 V. The DH signal is boosted to 6 V, so a high threshold for the lower transistor incurs no penalty.
The MOSFET transistors contain intrinsic diodes whose long recovery time can decrease efficiency. To prevent conduction in those diodes, four Schottky diodes, D1-D4, have been added across the four MOSFETs. Small 0.5-A packages are sufficient, because the Schottky diodes conduct only for brief intervals.
Each side of the H-bridge drives a low-pass filter, which consists of a 10-mH inductor followed by a 10-mF ceramic capacitor that feeds the TEC. A supplementary 10-mF across the TEC eliminates the possibility of residual spikes in differential mode. It's not necessary to oversize the inductors. A 20-mohm shunt is inserted for measuring the TEC current. Rough filtering (7.5 ohms and 1 mF) provides a clean 20-mV/A signal by eliminating a good part of the switching frequency ripple.
The signal is differentially amplified 32 times by the amplifier, which needs rail-to-rail input capability, because the input common-mode voltage ranges between ground and the supply voltage. An offset of 1.1 V is added to allow a single-polarity power supply. The output then indicates 1.1 V for the zero-current condition and deviates to either side with a sensitivity of 635 mV/A flowing into the TEC. Additional filtering eliminates the residual high-frequency ripple.
PWM controller. The heart of the regulator is the PWM controller (see Figure 3). This circuit works very well at power-supply voltages as low as 3.15 V when associated with low-threshold external MOSFETs. The controller used in the device discussed here, though not designed primarily for bidirectional current regulation, has been modified for that purpose. It provides two complementary signals, DH and DL, which in this case switch at 200 kHz. A 60-nsec dead time is automatically inserted to avoid cross-conduction between the external transistors, but the SKIP pin should be tied to supply voltage VCC to ensure complementariness between DH and DL.
The floating-gate driver output DH supplies enough voltage to saturate the n-channel upper-leg device. It is biased by the boost diode D5, which charges the 1-mF reservoir capacitor C1 whenever DL is active. The duty cycle doesn't exceed 96%, so C1 is always charged. At the other extreme, the duty cycle can reach 0%. The circuit exploits this asymmetry by reserving the low-duty-cycle region for cooling, which demands the most power.
The current-mode controller is able to sense current into the load, but it is not designed to accept the bidirectional currents present in this application. Accordingly, this function is disabled by connecting pins CSL and CSH to the 1% internal reference (REF). This reference is available as a convenience, especially when the voltage bridge is supplied by a more accurate source. For clean startups, the shutdown (SHDN) pin should be driven by an external source or generated locally by a reset circuit.
A precision amplifier performs the node-summation function. The PWM controller's under- and over-voltage protection triggers if the feedback (FB) input makes an excursion beyond the range of its normal working voltage. The voltage range is clipped by diode D6 and the resistive network driving the FB node. Amplifier B introduces a compensation pole for the internal loop, and the 100-nF capacitor ensures that unity gain is attained well before the frequency at which the LC H-bridge filter introduces too much phase lag.
Bridge amplifier. The bridge-amplifier function is implemented by two precision amplifiers in-series, configured in the inverting mode. Resist the temptation of fitting these amplifiers in the same package, because they are likely to present high gain at high frequency. They should be separated by a prudent distance to eliminate any possible coupling. (Lower- accuracy controllers can easily do this job with just one amplifier.)
Figure 4 gives an idea of the compensation network required in most cases. High-accuracy temperature regulators operate at high open-loop gain to ensure precision, but that condition may affect stability. A careful evaluation of each pole within the closed loop must include the worst-case variation of all parameters (component values, etc.) affecting the pole. The dominant pole in this case (caused by the thermal mass of the system) has a time constant of about 40 sec ±10 sec. The next highest pole is caused by the NTC sensor. An NTC time constant can range from 100 msec to 3 sec, depending on the model.
Those two low-pass-filter poles (dominant and NTC) are in-series, and if not compensated, then will obviously compromise the closed-loop stability. The third important pole is associated with the time constant of the intrinsic loop, which should be as small as possible to obtain good supply-voltage rejection. Because the other following poles (due to the LC H-bridge filters and differential filters) are not much higher in frequency, it is wise to set unity gain for the external loop well under the third pole. An actual model can be even more complicated, because coupling can occur within the thermal head. Some coupling is inevitable, because there is no way to isolate thermal blocks as we do with resistive electronic paths.
Therefore, it is wise to design for the most difficult case imaginable, then simplify the design if possible. Some tips can be suggested. We notice that a fairly high value was chosen for R36, which ensures that the bridge is not loaded even at high frequency. Capacitor C32 ensures a welcome excess of gain at very low frequencies. Resistor R38 is then chosen so the C32/R38 pole coincides with the 40-sec thermal pole. To be effective, C32 must have very high insulation resistance at the highest operating temperature.
Metallized polyester (PET) capacitors can provide time constants as high as 5,000 sec at 20°C, but that value drops rapidly with increasing temperature. Polyethylene naphtalate (PEN) is a better material at high temperatures. Obviously, severe precautions must be taken when designing the printed circuit for such high-impedance components. Large intervals must be provided between high-impedance tracks and insulation such as varnish added to protect against possible condensation.
The bridge is supplied by an accurate reference voltage of 2.75 V, which also biases all amplifiers in the system. The device's precision reference exhibits a temperature coefficient of 20 ppm/°C maximum. For every temperature error, the module output (1.1 V nominal) demands some positive or negative current. The resistive divider R43/R42 allows the TEC to be protected by setting different maximum limits for the heating demand and cooling demand. The minimum output voltage will be near zero (thanks to U4's rail-to-rail capability), and the resistor values shown produce a maximum voltage around 2 V. With a sensitivity of 635 mV/A in the circuit, the maximum currents are 1.65 A for cooling and 1.4 A for heating.
Efficiency and temperature stability. The temperature controller is associated with an equivalent 1.5-ohm TEC within an ultra-miniature module. The maximum heating and cooling currents are limited to 1.6 and 1.4 A, respectively, which corresponds to 3.84 W of available cooling power. Severe compromises have been made to allow an acceptable efficiency within the tiny space available. Component heights should be no greater than 3.5 mm. Copper thickness in the eight-layer printed circuit is only 17 microns, which is fairly resistive. Moreover, component placements in obstructed areas force the use of long and lossy connections.
Despite these limitations, the circuit has respectable efficiency (see Figure 5). The curves are blue for cooling, red for heating, and yellow for a linear controller. To account for electrical effects only, the measurements were made with an actual resistance of 1.71 ohms instead of the TEC. We notice that switch-mode (versus linear) control provides much greater efficiency throughout the current range. As a result, the switching module is able to provide greater cooling or accept a more elevated ambient temperature for a given level of input power (similarly for heating).
Power for cooling and heating is similar, but the cooling efficiency is somewhat better, thanks to low RDSon in the p-channel MOSFET Q1 (see Figure 2). Between 0.8 and 1.6 A, the efficiencies are almost flat at 84%, which means the switching and biasing losses are low and have little effect in that region of current. At high-level currents, the controller's equivalent output resistance (about 330 mohms) is mostly due to the printed circuit and connectors.
The module was tested over the range -5° to +70°C. Because the laser-diode wavelength is sensitive to temperature in a known and accurate manner, it was possible to verify stability for the diode temperature within ±0.1°C over the operating-temperature range.
Optimization tips. Printed-circuit traces can account for a large portion of the loss at high current. If possible, use copper layers of 35-micron thickness or greater. Also when possible, use multiple layers to create parallel tracks for the heavy current and connect the sister-tracks with plenty of vias to reduce parasitic resistance. Use the same trick for ground planes, which can benefit from the use of all available layers in the unused areas.
The most sensitive circuit is the bridge amplifier. Avoid common-mode voltage errors due to voltage drops in the ground plane. It's easy to accumulate millivolt errors at high current levels: At 1.6 A, the error should be below 2.7 mV (0.1°C). Provide a large copper surface for cooling the MOSFET power transistors, because their RDSon increases very fast with temperature. Finally, use strong ferrite-bead filtering to eliminate reverse contamination of the input supply. Contrary to the behavior of a classic inductor, the dissipation (loss) in a bead is low at DC but conveniently increases with frequency.
Thierry Rahban is a senior corporate applications engineer with Maxim Integrated Products (Sunnyvale, CA).